Methods and circuits for measuring mutual and self capacitance

Description

CROSS REFERENCE TO RELATED APPLICATION

This application is a continuation of U.S. patent application Ser. No. 13/664,421, filed Oct. 31, 2012, now U.S. Pat. No. 8,570,052, issued Nov. 29, 2013, which is a continuation of U.S. patent application Ser. No. 12/606,147, filed Oct. 26, 2009, now U.S. Pat. No. 8,319,505, issued Nov. 27, 2012, which claims the benefit of U.S. Provisional Patent Application No. 61/108,450, filed Oct. 24, 2008, and which is a continuation-in-part of U.S. patent application Ser. No. 12/395,462, filed Feb. 27, 2009, now U.S. Pat. No. 8,358,142, issued Jan. 22, 2013, which claims the benefit of U.S. Provisional Patent Application No. 61/067,539, filed Feb. 27, 2008, all of which are incorporated by reference herein in their entirety.

TECHNICAL FIELD

The present disclosure relates generally to touch sensors and, more particularly, to capacitive touch sensors.

BACKGROUND

Capacitive touch sensors may be used to replace mechanical buttons, knobs and other similar mechanical user interface controls. The use of a capacitive sensor allows for the elimination of complicated mechanical switches and buttons, providing reliable operation under harsh conditions. In addition, capacitive sensors are widely used in modern customer applications, providing new user interface options in existing products.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments are illustrated by way of example and are not intended to be limited by the figures of the accompanying drawings, in which like references indicate similar elements and in which:

FIG. 1 illustrates two electrodes situated close to each other, according to one embodiment;

FIG. 2 illustrates one embodiment of a self-capacitance circuit that uses a charge accumulation technique;

FIG. 3 illustrates a block diagram of an apparatus for measuring mutual or self capacitance, according to one embodiment;

FIG. 4A illustrates one embodiment of a capacitance to current sink converter having an integration capacitor coupled to ground;

FIG. 4B illustrates one embodiment of a capacitance to current sink converter having an integration capacitor coupled to a high voltage supply potential;

FIG. 5A illustrates one embodiment of a capacitance to current source converter having an integration capacitor coupled to ground;

FIG. 5B illustrates one embodiment of a capacitance to current source converter having an integration capacitor coupled to a high voltage supply potential;

FIG. 6A illustrates a first phase of a converter operation, according to one embodiment;

FIG. 6B illustrates a second phase of a converter operation, according to one embodiment;

FIG. 7A illustrates one embodiment of a capacitance to current sink converter used for mutual capacitance measurement, having an integration capacitor coupled to ground;

FIG. 7B illustrates one embodiment of a capacitance to current sink converter used for mutual capacitance measurement, having an integration capacitor coupled to V_DD;

FIG. 8A illustrates one embodiment of a capacitance to current source converter having an integration capacitor coupled to ground;

FIG. 8B illustrates one embodiment of a capacitance to current source converter having an integration capacitor coupled to a high voltage supply potential;

FIG. 9A illustrates one embodiment of a capacitance to current sink converter used for self capacitance measurement, having an integration capacitor coupled to ground;

FIG. 9B illustrates one embodiment of a capacitance to current sink converter used for self capacitance measurement, having an integration capacitor coupled to a high voltage supply potential;

FIG. 10A illustrates one embodiment of a capacitance to current source converter used for self capacitance measurement, having an integration capacitor coupled to a high voltage supply potential;

FIG. 10B illustrates one embodiment of a capacitance to current source converter used for self capacitance measurement, having an integration capacitor coupled to ground;

FIG. 11 illustrates one embodiment of an interval timer method for capacitance measurement;

FIG. 12 illustrates one embodiment of a resettable current integrator with an operation amplifier and an analog-to-digital converter (ADC);

FIG. 13 illustrates one embodiment of a current-to-voltage converter built around an operational amplifier;

FIG. 14 illustrates one embodiment of a capacitance to current converter with a conversion circuit;

FIG. 15 illustrates one embodiment of a capacitance to current converter with a low-pass filter;

FIG. 16 illustrates one embodiment of a sigma-delta modulator configured as a capacitance to duty cycle converter;

FIG. 17 illustrates one embodiment of a low pass filter with a differential analog to digital converter;

FIG. 18A illustrates base capacitance current compensation using a resistor as a current sink in a capacitance to current converter, according to one embodiment;

FIG. 18B illustrates base capacitance current compensation using a resistor for a current source in a capacitance to current converter, according to one embodiment;

FIG. 19A illustrates base capacitance current compensation using a current source as a current sink in a capacitance to current converter, according to one embodiment;

FIG. 19B illustrates base capacitance current compensation using a current source in a capacitance to current converter, according to one embodiment;

FIG. 20A illustrates using a current mirror with a voltage conversion system, according to one embodiment;

FIG. 20B illustrates using a current mirror with a current conversion system, according to one embodiment;

FIG. 20C illustrates one embodiment of a current mirror using a bipolar process technology; and

FIG. 21 illustrates one embodiment of a capacitance measurement circuit in a multi-touch touchpad system.

FIG. 22 illustrates one embodiment of a capacitance to current converter with a conversion circuit comprising a current mirror, an integration circuit and a timer.

DETAILED DESCRIPTION

In the following description, for purposes of explanation, numerous specific details are set forth in order to provide a thorough understanding of embodiments of the present invention. It will be evident, however, to one skilled in the art that embodiments of the present invention may be practiced without these specific details. In other instances, well-known circuits, structures, and techniques are not shown in detail or are shown in block diagram form in order to avoid unnecessarily obscuring an understanding of this description.

Reference in the description to “one embodiment” or “an embodiment” means that a particular feature, structure, or characteristic described in connection with the embodiment is included in at least one embodiment of the invention. The appearances of the phrase “in one embodiment” in various places in the specification do not necessarily all refer to the same embodiment. Like reference numerals denote like references elements throughout.

A capacitive sensor may be characterized by a base capacitance that includes a self capacitance component and a mutual capacitance component. Since the values of these capacitance components affect the operation of the capacitive touch sensor and may vary from one capacitive sensor to another, a capacitive sensing circuit may benefit from the capability of independently measuring the self and mutual capacitances of a capacitive sensor.

Apparatus for and methods of measuring mutual and self capacitance in a capacitive touch sensor are described. The apparatus and methods described herein may be used in capacitive touch detection systems such as, for example, capacitive touch screens and, in particular, with capacitive touch screens having multiple simultaneous touch detection capabilities. Alternatively, the apparatus and methods described herein may be used with single touch detection systems or other types of capacitive touch systems.

The capacitance measurement circuits described herein may be used for touch detection in single electrode systems, transmit/receive (TX-RX) systems, or in combined TX-RX and single electrode systems. The TX-RX systems can use the mutual capacitance change detection, and single electrode systems can use the self capacitance change detection. In some embodiments, additional multiplexers can be added for multiple electrode scanning. In other embodiments additional capacitance-to-current converters may be added to allow parallel scanning of multiple sensor electrodes. The capacitance measurement circuits described herein may be used in various applications including, for example, single button applications, multiple buttons applications, linear and radial sliders, dual dimension touchpads and touchscreens, and multi-touch touchpad and touchscreen applications. Multi-touch touchpad and touchscreen systems are composed of a matrix of RX and TX electrodes, where the presence (e.g., touch) of a finger (or other conductive object) is detected as a decrease in the mutual capacitance at the intersection of the TX-RX electrodes.

Embodiments of the present invention allow for measurement of two or more ‘electrodes’ mutual and self capacitance separately. Capacitance measurement can be performed with a single pair of electrodes or with the use of a multiple electrode system. Two electrodes situated close to each other are shown at FIG. 1, where C_e1 101 and C_e2 102 are electrode self capacitances, and C_m 103 is the mutual capacitance between the two electrodes E₁ 104 and E₂ 105.

There are various circuit implementations that may be used for performing capacitance measurement. FIG. 2 illustrates a self-capacitance circuit 200 that uses a charge accumulation technique to measure the capacitance C_X 204. A charge accumulation technique operates in the following way: initially the integration capacitor 203 is reset by turning on a reset signal for some time which sets switch 205 such that both ends of integration capacitor 203 are grounded. After reset, the switches 201 and 202 start operation in the two non-overlapping phases, wherein switch 201 when closed accumulates charge onto C_x and wherein switch 202 when closed allows that charged to be integrated onto C_int 203. As more charge is integrated onto integration capacitor C_int 203, the voltage on C_int 203 starts increasing. The sensing capacitance may be determined by the number of switching cycles used to get the integrator capacitor voltage to some threshold value, such as V_DD/2 or a bandgap voltage (V_BG).

With such a charge accumulation technique, the voltage on the integration capacitance rises exponentially with respect to time (said time can be measured by a count of the number of cycle it takes to reach the threshold value). This relationship can be linearized for measurement methods where capacitance is calculated as a function of integration capacitor voltage after a predefined number of cycles. Also, the mutual capacitance measurement scheme has some sensitivity to the sensor self capacitance, which decreases the measurement accuracy.

FIG. 3 illustrates a block diagram of a capacitance measurement circuit 300 for measuring mutual or self capacitance, according to one embodiment of the present invention. The apparatus illustrated in FIG. 3 can be used for separately measuring mutual or self capacitances of a capacitance sensor. In order to measure a mutual capacitance, the C_e1, C_e2 (of FIG. 1) capacitance influence should be excluded. This can be accomplished by charging and discharging the C_e2 electrode from a low-impedance voltage source and keeping the voltage of the C_e1 electrode close to constant to minimize the influence of its charge-discharge current. In order to measure the self-capacitance (of C_e1 or C_e2) the voltage change across C_m should be kept to zero to minimize the influence of this capacitance on the measurement results.

The capacitance measurement circuit 300 can be separated into two parts: the switching capacitor front-end capacitance-to-current converter 301, and the back-end current-to-digital value converter 302, as illustrated in FIG. 3. In the following description, the front-end and back-end circuits are described separately. A switching capacitor front-end converts the sensing capacitance to current pulses. The back-end system averages the current and converts it into readable digital values. The circuits described herein are based on a switching capacitor technique in capacitance-to-current converter circuits.

FIGS. 4A, 4B, 5A and 5B show different embodiments for a capacitance to current converter for mutual capacitance measurement. In the referenced figures, a voltage buffer 401 resides between the integration capacitor C_int 406 and the switches 402, 404 connecting to the mutual electrodes of the capacitance-to-current. It should be noted that the integration capacitor C_int 406 is considered as part of the current measurement system and shown here for ease of explanation. The integration capacitor 406 can be connected between the converter output and a fixed potential net, for example, GND and V_CC, as illustrated in FIGS. 4A, 4B, 5A and 5B, respectively.

The operation of the circuit may be described in several stages, which are repeated in cycle. Table 1 contains the switching sequence of switches for the circuits shown in FIGS. 4A and 4B.

TABLE 1
Switching sequence of switches shown in FIGS. 4A and 4B.

	Switch	Switch	Switch	Switch
Stage	402	403	404	405	UCint, UCe1, UCe2, UCm

1	OFF	OFF	OFF	OFF	UCint = U0
2	ON	OFF	ON	OFF	UCm = 0,
					UCe1 = UCe2 = UCint = Ubuf
3	OFF	OFF	OFF	OFF	UCm = 0,
					UCe1 = UCe2 = UCint
4	OFF	ON	OFF	ON	UCm = UCint = UCe1 ,
					UCe2 = 0
5	OFF	OFF	OFF	OFF	UCm = UCe1 , UCe2 = 0

Table 2 contains the switching sequence of switches for the circuits shown in FIGS. 5A and 5B.

TABLE 2
Switching sequence of switches shown in FIGS. 5A and 5B.

	Switch	Switch	Switch	Switch
Stage	402	403	404	405	UCint, UCe1, UCe2, UCm
1	OFF	OFF	OFF	OFF	UCint = U0
2	OFF	ON	ON	OFF	UCm = Ubuf = UCint = UCe1
3	OFF	OFF	OFF	OFF	UCm = UCint = UCe1
4	ON	OFF	OFF	ON	UCm = 0, UCe1 = UCint,
					UCe2 = UCint
5	OFF	OFF	OFF	OFF	UCm = 0, UCe1 = UCint,
					UCe2 = UCint

The stages from 2 to 5 are performed in cycles. In effect, the circuits shown in FIGS. 4A and 4B may act as current sinks, and the circuits shown in FIGS. 5A and 5B may act as current sources in the respective embodiment. The integration capacitor C_int 406 is external to the capacitance-t0-current converter and is not part of the current measurement circuit.

FIGS. 6A and 6B illustrates one embodiment of the operation phases for the circuits shown in FIGS. 4A and 4B, respectively. During the first phase (FIG. 6A), both ends of the C_m 103 are connected to voltage buffer 401. During the second phase (FIG. 6B), the left C_m terminal is grounded and the right terminal is connected to the integration capacitor C_m, 406.

For both circuits, an averaged absolute current sink/source (I_S) value can be calculated by Equation 1:
I_S=f_sw·U_Cint·C_m  (1)
where, f_sw is the switching frequency of phases 2-5 repeating. It should be noted that the capacitance of C_e2 electrode 102 is shunted by switch 402 or 403 in each operation phase and does not have an impact on the output current. The capacitance of the C_e1 electrode 101 has a potential equal to U_Cint during both charge transfer stages and is not recharged between different operation phases. Therefore, the output current is determined by the value of C_m 103 and the potential applied across it.

A special case of the capacitance-to-current converter operation is now considered, when it is loaded by stand-alone integration capacitor C_int 406. In this case, the relationship between the voltage change on U_Cint and the cycles count N has a nonlinear exponential character, as expressed in Equation 2:

U C ⁢ ⁢ int N = U Cint 0 · ( 1 - C m C int ) N ( U C ⁢ ⁢ int N ≈ U Cint 0 · ⅇ - N ⁢ C m C int ) ( 2 )
where, N is the quantity of conversion cycles and U_Cint⁰ is the voltage on the integration capacitor 406 at the initial time.

The exponential character of this dependence is caused by the positive voltage feedback via buffer 401: increasing voltage on the integration capacitor 406 (when the capacitance-to-current converter is configured as a current source) causes a larger charge quantum being moved in each phase and an increase in the speed of the integration capacitor 406 voltage rising. The current measurement circuit may not keep a voltage on the integration capacitor 406 constant in this embodiment.

The circuit embodiments illustrated in FIGS. 7A, 7B, 8A, and 8B may be used to keep voltage on C_int 406 constant. The difference between the circuit embodiments illustrated in FIGS. 7A, 7B, 8A, and 8B, versus those illustrated in FIGS. 4A, 4B, 5A, and 5B, is that the left terminal of C_m 103 is connected to the fixed voltage source V_DD in FIGS. 7A, 7B, 8A and 8B. In FIGS. 7A, 7B, 8A and 8B the variable buffer output voltage of an analog buffer 701 is coupled to the right terminal of C_m 103 and in FIGS. 4A, 4B, 5A and 5B buffer 401 is coupled with the left terminal of C_m 103. Only the switch 702 connection is changed on the circuits illustrated in FIGS. 7A, 7B, 8A, and 8B.

The switching sequence of the switches illustrated in FIGS. 7A and 7B is shown in Table 3.

TABLE 3
Switching sequence of switches in FIGS. 7A and 7B.

	Switch	Switch	Switch	Switch
Stage	702	703	704	705	UCint, UCe1, UCe2, UCm
1	OFF	OFF	OFF	OFF	UCint = U0
2	ON	OFF	ON	OFF	−UCm = UVdd − UCint,
					UCe1 = UCint = Ubuf,
					UCe2 = UVdd
3	OFF	OFF	OFF	OFF	UCm = UVdd − UCint,
					UCe1 = UCint , UCe2 = UVdd
4	OFF	ON	OFF	ON	UCm = UCint = UCe1,
					UCe2 = 0
5	OFF	OFF	OFF	OFF	UCm = UCe1, UCe2 = 0

The switching sequence of the switches illustrated in FIGS. 8A and 8B is shown in Table 4.

TABLE 4
Switching sequence of switches in FIGS. 8a and 8b

	Switch	Switch	Switch	Switch
Stage	702	703	704	705	UCint, UCe1, UCe2, UCm
1	OFF	OFF	OFF	OFF	UCint = U0
2	OFF	ON	ON	OFF	UCm = Ubuf = UCint = UCe1
3	OFF	OFF	OFF	OFF	UCm = UCint = UCe1
4	ON	OFF	OFF	ON	−UCm = UVdd − UCint,
					UCe1 = UCint, UCe2 = UVdd
5	OFF	OFF	OFF	OFF	UCm = 0, UCe1 = UCint,
					UCe2 = UVdd

The stages from 2 to 5 are performed in cycles. As a result, the average current flowing out of the C_int 406 capacitor for the circuits on FIGS. 7A, 7B, 8A, and 8B may be calculated by Equation 3:
I_S=f_sw·U_Vdd·C_m  (3)
For the given values of f_sw and V_DD parameters, the output current (I_S) linearly depends on C_m and is proportional to f_sw and V_DD in this embodiment. The change of current direction is done by a change of the switches' operation phases. If the current measurement subsystem does not load the integration capacitor C_int 406, a voltage on this capacitor changes linearly with the number of cycles N, as expressed in Equation 4:

U C ⁢ ⁢ int N = U Vdd · ( 1 - N · C m C int ) ( 4 )

A similar Equation 5 is used for describing the circuits illustrated in FIGS. 8A and 8B:

U N = N · U Vdd · C m C int ( 5 )

The mutual capacitance circuit embodiments described previously may be used for self-capacitance measurement with minimal hardware changes by routing the buffer signal to the left-side switches. To do this, the switched voltages may be adjusted in such a way that the voltage change on the mutual capacitance C_m is equal to zero between different phases. In other circuit configurations, the voltage on C_e2 is kept constant but the voltage on C_m is varied. In the circuit embodiments illustrated in FIGS. 7A, 7B, 8A, and 8B, the voltage on C_e2 is varied and the voltage change on C_m is kept constant.

FIGS. 9A and 9B illustrate embodiments of a capacitance to current sink converter for self capacitance measurement. As previously noted, the integration capacitor C_int 406 is considered part of the current measurement system and is shown here for ease of explanation. The integration capacitor 406 can be connected between the converter output and any fixed potential net, for example, GND and V_CC, as illustrated in FIGS. 9A and 9B respectively. Alternatively, the integration capacitor 406 can be connected between the converter output and other fixed potentials.

The switching sequence of switches illustrated in the circuit of FIGS. 9A and 9B is shown in Table 5.

TABLE 5
Switching sequence of switches illustrated in FIGS. 9A, 9B.

	Switch	Switch	Switch	Switch
Stage	902	903	904	905	UCint, UCe1, UCe2, UCm
1	OFF	OFF	OFF	OFF	UCint = U0
2	OFF	ON	ON	OFF	UCe1 = UCe2 = 0,
					UCm = 0
3	OFF	OFF	OFF	OFF	UCe1 = UCe2 = 0,
					UCm = 0
4	ON	OFF	OFF	ON	Ue1 = UCint = UCe2,
					UCm = 0
5	OFF	OFF	OFF	OFF	Ue1 = UCint = UCe2,
					UCm = 0

FIGS. 10A and 10B illustrate embodiments of a capacitance to current source converter for self capacitance measurement. As previously noted, the integration capacitor C_int 406 is considered part of the current measurement system and is shown here for ease of explanation. The integration capacitor 406 can be connected between the converter output and any fixed potential net, for example, GND and V_CC, as illustrated in FIGS. 10A and 10B respectively. Alternatively, the integration capacitor 406 can be connected between the converter output and other fixed potentials.

The switching sequence of switches in FIGS. 10A and 10B is shown in Table 6.

TABLE 6
Switching sequence of switches illustrated in FIGS. 10A, 10B.

	Switch	Switch	Switch	Switch
Stage	902	903	904	905	UCint, UCe1, UCe2, UCm
1	OFF	OFF	OFF	OFF	UCint = U0
2	ON	OFF	ON	OFF	UCe1 = UCe2 = UVdd,
					UCm = 0
3	OFF	OFF	OFF	OFF	UCe1 = UCe2 = UVdd,,
					UCm = 0
4	OFF	ON	OFF	ON	Ue1 = UCint = UCe2,
					UCm = 0
5	OFF	OFF	OFF	OFF	Ue1 = UCint = UCe2,
					UCm = 0

Stages 2 through 5 are performed in cycles. As a result, the average current flowing out of capacitor C_int for the circuits illustrated in FIGS. 9A and 9B may be described by Equation 6:
I_S=f_sw·U_Cint·C_e1  (6)

The average current flowing into C_int capacitor for the circuits illustrated in FIGS. 10A and 10B may be described by Equation 7:
I_S=f_sw·(U_Vdd−U_Cint)·C_e1  (7)

The potential difference on electrode capacitor C_m 103 is equal to approximately zero during the stages of charge transfer and does not have an impact on the measurement in this embodiment. The C_e2 electrode 102 capacitance is switched off by switches 902 and 904 during the stages of operation. In this case, the relationship between the voltage change on U_Cint and the cycle count N has a nonlinear exponential character for the circuits illustrated in FIGS. 9A and 9B, in accord with Equation 8:

U C ⁢ ⁢ int N = U Cint 0 · ( 1 - C e ⁢ ⁢ 1 C int ) N ( 8 )

Equation 9 similarly describes the circuits illustrated in FIGS. 10A and 10B:

U C ⁢ ⁢ int N = U Vdd · ( 1 - ⅇ - N · C e ⁢ ⁢ 1 C int ) ( 9 )

Various alternative variants of the conversion circuits described above may be used. Alternative conversion circuits include integration circuits such as time measurement of the integration capacitor voltage threshold crossing, current integrations with an operational amplifier as a current integrator. Alternative conversion circuits include analog-to-digital circuits such as a current-to-voltage conversion using an operational amplifier and an ADC voltage measurement device or sigma-delta modulation circuits.

FIG. 11 illustrates an interval timer method for capacitance measurement. In the circuit of FIG. 11, the integrator consists of a capacitor 406. The circuit of FIG. 11 operates in the following way. Initially, the voltage of integration capacitor 406 is set to I_init by turning on, for some time period, a switch 1102. The comparator 1101 is used as threshold circuit and compares the voltage on the integration capacitor 406 with a reference voltage U_ref. The capacitance is measured by the time measurement circuit 1103 as the time elapsed (in the cycles count) until the comparator 1101 is triggered. The time is inversely proportional to the capacitance-to-current converter current in this embodiment. It should be noted that for switching capacitor current sink schemes, an integrator initial voltage (U_init) is set higher than the threshold voltage (U_ref). For the current source schemes, the integrator initial voltage is lower than threshold voltage U_ref.

For more accurate current conversion, circuits based on current-to-voltage converters and current integrators may be used, as illustrated in FIGS. 12 and 13. FIG. 12 illustrates one embodiment of a resettable current integrator (where integration capacitor 1203 can be reset using switch 1204) with an operational amplifier 1201 with reference input voltage V_ref and an analog-to-digital converter (ADC) 1202 coupled to the output of operational amplifier 1201. The ADC 1202 is used for integrator voltage measurement after the completion of a predefined number of integration cycles. Capacitance on the is converted to current by capacitance-to-current converter 301 and C_int 1203 is charged with the output current to Vref on the operational amplifier 1201. The output of operational amplifier 1201 coupled to the opposite side of C_int 1203 is then measured by ADC 1202 to calculate the value of the capacitance in the capacitance-to-current converter 301.

FIG. 13 illustrates one embodiment of a current-to-voltage converter built around an operational amplifier 1301. The converter of FIG. 13 also functions as a low pass filter (LPF) due to the presence of the filter capacitor C_filt 1302 in the amplified feedback path. The output voltage U_S is proportional to the input current I_S. The circuit of FIG. 13 operates continuously such that ADC conversion can be started any time after transient signals have stabilized. It should be noted that the buffer input inside the capacitance-to-code converter can be connected to the V_ref net for the circuits illustrated in FIGS. 12 and 13, taking into account that both operational amplifier inputs have approximately the same potential. The schematic diagram of such a circuit configuration is illustrated in FIG. 14, where the input of voltage buffer 1401 is connected to the V_ref net.

In an alternative embodiment, when the V_ref voltage source has an acceptable low output resistance, then the voltage buffer 1401 may be eliminated from the circuits illustrated herein. As an example, the circuit from FIG. 4 composed of the measurement circuit of FIG. 13 is illustrated in FIG. 15. Accordingly, FIG. 15 is an example illustration of a capacitance-to-code converter with a low-pass filter that can be implemented without a voltage buffer 1401 coupled to the reference voltage source V_ref. In one embodiment, the reference voltage V_ref used to supply the switches in the capacitance-to-current converter is selected to be as close to V_DD as possible (limited by the working range of the operational amplifier 1301), to minimize the current flow out of C_e2 102 relative to the current flowing through C_m 103. In alternative embodiments, the switches in the converters can be supplied with other known voltages such as, for example, V_DD.

The sigma-delta modulator circuits can be effectively used for the current-to-code conversion. An advantage of the sigma-delta modulator circuits is their integrative nature. FIG. 16 illustrates one possible example of a modulator implementation for a first order modulator. It should be noted that higher order modulator circuits can be used as well. The sigma-delta modulator of FIG. 16 converts the current I_S to a code in output bitstream 1601. The current I_S discharges modulation capacitor C_mod 1602 until the voltage at C_mod 1602 falls below V_ref, at which point comparator 1603 asserts its output to latch 1604, which outputs bits synchronously with a clock signal provided by clock 1605. The latch 1604 then closes switch 1606 to recharge C_mod 1602 at a rate faster than it is being discharged by current I_S. When the voltage at C_mod 1602 rises above V_ref, comparator 1603 de-asserts its output to latch 1604, which then opens synchronously to open switch 1606, allowing current I_S to again discharge C_mod 1602. This toggling of state of latch 1604 provides the bitstream 1601 which may then be evaluated to provide a digital-code equivalent of the measured capacitance.

In one embodiment, the capacitance measurement circuit embodiments described previously may be used in touch sensitive devices. With such devices, a small capacitance change should be detected over the presence of large base capacitance. Such sensors have two components of capacitance, described in Equation 10:
C_S=C_Sconst+C_Stouch  (10)
where C_Sconst is the capacitance of a sensor when touch is absent, and C_Stouch is the change in capacitance caused by an input, such as a finger touch. The informative part of the sensor capacitance C_S is the C_Stouch component. In order to increase the resolution of the sensor, the particular compensation of the current generated by the C_Sconst capacitance can be used. There are several possible implementations of this technique. In one embodiment, an ADC 1701 with differential inputs may be used as illustrated in FIG. 17. In the circuit of FIG. 17, the U_comp voltage is supplied to the second input of ADC 1701.

Alternative embodiments provide base-capacitance current compensation using a programmable current source or a resistor, as illustrated in FIGS. 18A, 18B, 19A, and 19B. More specifically, FIG. 18A illustrates base-capacitance current compensation using a resistor R_bias 1801 as a current source in a capacitance-to-current converter, according to one embodiment. FIG. 18B illustrates base capacitance current compensation using a resistor R_bias 1811 as a current sink in a capacitance-to-current converter, according to one embodiment. Current source 1901 and current source 1911 add and subtract current to and from the capacitance-to-current conversion circuits current onto C_int 1302, respectively. FIG. 19A illustrates base-capacitance current compensation using a current source 1901 as a current sink in a capacitance-to-current converter, according to one embodiment. FIG. 19B illustrates base-capacitance current compensation using a current source 1911 in a capacitance-to-current converter, according to one embodiment. The R_bias 1801 and R_bias 1811 add and subtract current to and from the capacitance-to-current conversion circuits current onto C_int 1302, respectively.

FIGS. 20A, 20B, and 20C illustrate using a current mirror 2001 in the conversion circuits. FIG. 20A shows an example of a circuit for current-to-voltage conversion using a low-pass filter, formed by the combination of load resistance R_L 2002 and filter capacitor C_filt 2003. The current to voltage conversion is carrying out on the output of capacitance-to-current converter 301. A filter output voltage can be measured using an ADC, such as that shown FIG. 12 (element 1202). FIG. 20B illustrates a current-to-current conversion circuit. A current (I_SM) is sourced to the integration capacitor C_int 2013 from current mirror 2001 which is coupled to and mirrors the current (I_S) of capacitance-to-current converter 301. The different circuits can be used for integration capacitor current measurement. In one embodiment, a current can be measured using a threshold comparator and a timer (not shown). In another embodiment, an integration capacitor voltage is measured using an ADC after running a capacitance-to-current operation within a predefined amount of time. The current mirror has low input impedance, which allows keeping a current mirror input pin voltage close to a constant voltage (e.g., V_DD). This embodiment may optimize the operating conditions of the capacitance-to-current conversion circuit, allowing the use of a voltage buffer with a smaller slew rate and reduced current consumption. Also, the current mirror serves as a current amplifier or attenuator, boosting or reducing the converter current by a factor of N. Many implementations of the current mirror circuit are possible, and one implementation using bipolar transistors is shown at FIG. 20C (element 2001). Equivalent circuits may be made with MOS, CMOS, and other circuit techniques.

FIG. 21 illustrates a simplified schematic of a multi-touch touchpad system. The multi-touch touchpad system 2100 is composed of a dual dimension array (matrix) of electrodes 2101 coupled to column and row signal multiplexers 2102 and 2103. Multiplexor control 2107, clock source 2104, drive switches 2105 couple electrodes matrix 2101 to capacitance-to-current converter 301. Capacitance-to-current converter 301 is coupled to current-to-code converter 302. Current-to-code converter 302 is coupled to touch coordinate estimation system 2106. The electrodes matrix 2101 can be fabricated from any conductive material, as copper, conductive ink, Indium Tin Oxide (ITO), PEDOT, etc.

FIG. 22 illustrates one embodiment of the capacitance-to-current converter comprising a current mirror 2250 and integration circuit (C_int 2230, switch 2206 and CMP 2240) and a timer 2260.

Sensitivity and resolution of the capacitance measurement circuit are increased and the scan time is reduced with the addition of the compensating current Um generated by the programmable current supply IDAC 2210. The result summary current I_in is generated and given by:
I_in=I_s−I_IDAC  (11)

Current mirror 2250 mirrors I_in into I_int with a transmission efficiency K₁=−1. A reset switch on current integrator capacitor C_int, along with a timer 2260 and a comparator CMP 2240, digitize the current from current mirror 2250. Before measurement, the voltage on capacitor C_int 2230 is reset to a reset potential by reset switch 2206. During operation, charge integration on C_int 2230 increases the voltage across C_int 2230 as:

U C ⁢ ⁢ int = I int · t int C int ( 12 )

The voltage across C_int, U_Cint, reaches a threshold V_ref on the comparator CMP 2240 input, causing a hold on timer 2260 which is started at the beginning of the charge integration. The output of the timer is a digital representation of the non-compensated capacitance of C_m+C_F. (wherein C_F is the change in C_m caused by the presence of a finger or other conductive object). The full equation for the conversion of capacitance to current to counts is given by:

T x = U V ⁢ ⁢ ref K 1 ⁡ ( f sw · U Vdd · C m - I IDAC - f sw · U Vdd · C F ) · C int ( 13 )

This timer output signal may be used to operate reset switch 2206. In an alternate embodiment, integration capacitor reset switch 2206 may be operated by a state-machine or under control of a CPU executing instructions. Before measuring, the timer register's values are configured to initial values. The integration time may be sampled by a timer clock signal with a period of

T ref = 1 f ref ( 14 )

The result of the value of the clock frequency cycle number in one integration cycle of the measured current is given by:

N x = T x T ref ± 1 ( 15 )

In the foregoing specification, the invention has been described with reference to specific embodiments thereof. It will, however, be evident that various modifications and changes may be made thereto without departing from the broader spirit and scope of embodiments of the invention.